Forward converter transformer saturation prevention

ABSTRACT

A power converter in one aspect limits the magnetic flux in a transformer. A control circuit included in the power converter includes a pulse width modulator, a logic circuit and a saturation prevention circuit. The saturation prevention circuit asserts a first signal when a first integral value of the input voltage reaches a first threshold value and asserts a second signal after a delay time that begins when a difference between the first integral value and a second integral value of a reset voltage of the transformer falls to a second threshold value. The logic circuit turns off the switch when the first signal is asserted, and allows the switch to turn on and off in accordance with the pulse width modulator when the second signal is asserted.

REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/090,160, filed on Apr. 19, 2011, now pending, which is a continuationof U.S. patent application Ser. No. 12/950,783, filed on Nov. 19, 2010,now U.S. Pat. No. 7,952,898, which is a continuation of U.S. patentapplication Ser. No. 12/234,525, filed on Sep. 19, 2008, now U.S. Pat.No. 7,859,869. U.S. patent application Ser. No. 13/090,160 and U.S. Pat.Nos. 7,952,898 and 7,859,869 are hereby incorporated by reference.

BACKGROUND INFORMATION

1. Field of the Disclosure

The present invention relates generally to power supplies and, moreparticularly, the present invention relates to forward converters.

2. Background

AC-to-DC and DC-to-DC power supplies typically use a power conversiontopology commonly known in the art as a forward converter.

A forward converter may use either one or two active switches to applyan input voltage to the primary winding of a transformer. Thesingle-switch forward converter uses one active switch to apply an inputvoltage to the primary winding of a transformer. The two-switch forwardconverter uses two active switches to apply an input voltage to theprimary winding of a transformer. In each type of forward converter, asecondary winding on the transformer produces a scaled replica of thevoltage on the primary winding. The voltage on the secondary winding isrectified and filtered to become an output voltage.

In a power supply, the output voltage is normally regulated by a controlcircuit. The control circuit compares the output voltage to a desiredvalue. The control circuit turns the active switches on and off, andadjusts the time that the switches are on (and off) to keep the outputnear the desired value.

The choice of one or two switches in the design of a forward converteris heavily influenced by cost. The two-switch forward converter is oftenthe lowest cost configuration that meets the requirements of powersupplies for personal computers and similar applications.

Both the single switch configuration and the two switch configurationallow the magnetic flux of the transformer to reset (that is, return toa much lower value) when the active switches are off. Resetting themagnetic flux of the transformer prevents excess stored energy fromsaturating the transformer (which alters properties of the transformer).The reset is generally achieved by applying a reset voltage ofappropriate magnitude and duration to the primary winding when theactive switches are off.

It is often desirable to set the reset voltage to a higher value thanthe input voltage that appears on the primary winding when the switchesare on. A common low-cost technique to provide a suitable reset voltageuses a simple reset circuit to develop a substantially constant voltagethat is applied to the primary winding during the reset time of thetransformer. In a two-switch forward converter, the reset voltage is thesum of the input voltage and the voltage of the reset circuit. In asingle-switch forward converter, the reset voltage is the voltage of thereset circuit.

A difficulty with the technique that uses the simple reset circuit isthat the appropriate reset voltage can change suddenly when the controlcircuit responds to a change in the input voltage or to a change in theload on the power supply. Also, the simple reset circuit usually cannotrespond fast enough to transient events (such as the start-up and theshut-down of the power supply) to guarantee a proper reset of thetransformer.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the present invention aredescribed with reference to the following figures, wherein likereference numerals refer to like parts throughout the various viewsunless otherwise specified.

FIG. 1 is a schematic diagram that illustrates the salient features of atwo-switch forward converter with a control circuit that includes theinvention.

FIG. 2 is a schematic diagram that shows details of an example resetcircuit that may be used with the two-switch forward converter of FIG.1.

FIG. 3 is a block diagram that shows functional elements and signalsincluded in the control circuit of the two-switch forward converter ofFIG. 1.

FIG. 4 is a schematic diagram that illustrates one example of thefunctions included in the saturation prevention circuits included in thecontrol circuit of FIG. 3.

FIG. 5 is a timing diagram that shows signals of the saturationprevention circuits of FIG. 4 for a condition of normal operation thatdoes not activate the features that prevent saturation of thetransformer.

FIG. 6 is a timing diagram that shows the same signals as FIG. 5 for acondition that terminates a gate signal to prevent saturation of thetransformer.

FIG. 7 is a timing diagram that shows the same signals as FIG. 5 for acondition that delays a gate signal to prevent saturation thetransformer.

FIG. 8 is a schematic diagram that shows a portion of an integratedcircuit that produces signals included the saturation preventioncircuits of FIG. 4.

FIG. 9 is a schematic diagram that shows another portion of anintegrated circuit that produces signals included in the saturationprevention circuits block and the logic circuits block of FIG. 3.

FIG. 10 is a schematic diagram that shows yet another portion of anintegrated circuit that produces signals included in the logic circuitsblock of FIG. 3.

FIG. 11 is a timing diagram that shows signals from the portions of theintegrated circuit illustrated in FIG. 8, FIG. 9, and FIG. 10, for thesame conditions as the timing diagram of FIG. 5.

FIG. 12 is a timing diagram that shows signals from the portions of theintegrated circuit illustrated in FIG. 8, FIG. 9, and FIG. 10, for thesame conditions as the timing diagram of FIG. 6.

FIG. 13 is a timing diagram that shows signals from the portions of theintegrated circuit illustrated in FIG. 8, FIG. 9, and FIG. 10, for thesame conditions as the timing diagram of FIG. 7.

FIG. 14 is a flow diagram that illustrates a method to preventtransformer saturation in a forward converter.

FIG. 15 is a schematic diagram that illustrates the salient features ofa single-switch forward converter with a control circuit that includesthe invention.

FIG. 16 is a schematic diagram that illustrates another example of thefunctions included in the saturation prevention block included in thecontrol circuit of FIG. 3 for a single-switch forward converter.

DETAILED DESCRIPTION

Methods and apparatuses for implementing a proper reset of thetransformer with a relatively simple passive circuit to develop thereset voltage. In the following description, numerous specific detailsare set forth in order to provide a thorough understanding of thepresent invention. It will be apparent, however, to one having ordinaryskill in the art that the specific detail need not be employed topractice the present invention. In other instances, well-known materialsor methods have not been described in detail in order to avoid obscuringthe present invention.

Reference throughout this specification to “one embodiment”, “anembodiment”, “one example” or “an example” means that a particularfeature, structure or characteristic described in connection with theembodiment or example is included in at least one embodiment of thepresent invention. Thus, appearances of the phrases “in one embodiment”,“in an embodiment”, “one example” or “an example” in various placesthroughout this specification are not necessarily all referring to thesame embodiment or example. Furthermore, the particular features,structures or characteristics may be combined in any suitablecombinations and/or subcombinations in one or more embodiments orexamples. In addition, it is appreciated that the figures providedherewith are for explanation purposes to persons ordinarily skilled inthe art and that the drawings are not necessarily drawn to scale.

FIG. 1 is an illustration of an example two-switch forward converter 100in accordance with the present invention. The two-switch forwardconverter uses two active switches, S1 104 and S2 116, with two passiveswitches, D1 120 and D2 106 in a configuration that produces a voltageV_(p) 110 on a primary winding 112 of a transformer T1 114 from an inputvoltage V_(IN) 102. In the example of FIG. 1, the input voltage V_(IN)102 has a negative terminal that is common with an input return 108.

Active switch S1 104 is often referred to as a high side switch becauseit has one terminal common with the positive terminal of the inputvoltage V_(IN) 102. Active switch S2 116 is often referred to as alow-side switch because it has one terminal common with the input return108. Similarly, passive switch D1 120 is may be referred as a high-sideswitch and passive switch D2 106 may be referred to as a low-sideswitch.

A secondary winding 134 of the transformer T1 114 produces a voltageproportional to the primary voltage V_(P) 110. An output diode 136rectifies the voltage at the secondary winding 134. A freewheeling diode138, an output inductor L1 140, and an output capacitor C1 142 filterthe rectified voltage from the secondary winding 134 to produce anoutput voltage V_(O) at a load 144. In the example of FIG. 1, thenegative terminal of the capacitor C1 142 with output voltage V_(O) iscommon with an output return 152.

The secondary winding 134 of the transformer T1 114 typically isgalvanically isolated from the primary winding 112. That is, a DCvoltage between the primary return 108 and the secondary return 152normally produces substantially zero current between the primary return108 and the secondary return 152.

A difference between an active switch and a passive switch is that theactive switch receives a control signal that opens and closes the switchwhereas a passive switch does not receive a control signal. An openswitch does not normally conduct current. A closed switch may conductcurrent. Active switches typically have one or more control terminalsthat determine whether or not two other terminals of the active switchmay conduct current. In the example of FIG. 1, a gate signal 130 opensand closes active switches S1 104 and S2 116. In practice, switches S1104 and S2 116 are typically semiconductor devices such as for examplemetal oxide semiconductor field effect transistors (MOSFETs), or forexample bipolar junction transistors (BJTs), or for example insulatedgate bipolar transistors (IGBTs).

Passive switches generally have only two terminals. Typically, thevoltage between the terminals determines whether a passive switch isopen or closed. A diode is sometimes considered a passive switch, sinceit conducts current when the voltage between its two terminals has onepolarity (anode positive with respect to cathode), and it substantiallyblocks current when the voltage between the terminals has the oppositepolarity (anode negative with respect to cathode). The passive switchesD1 120 and D2 106 in the example of FIG. 1 are PN junction diodes.

In the two-switch forward converter of FIG. 1, a magnetic flux increasesin the transformer T1 114 when the active switches S1 104 and S2 116 areturned on, and the magnetic flux decreases in the transformer T1 114when the active switches S1 104 and S2 116 turn off. The magnetic fluxis associated with a magnetizing current that enters the windings of thetransformer when the active switches S1 104 and S2 116 are on. When theactive switches S1 104 and S2 116 turn off, the magnetizing currentleaves the primary of the transformer T1 114 through the passiveswitches D1 120 and D2 106. A reset circuit 118 produces a reset circuitvoltage V_(RC) 150 from the magnetizing current that leaves thetransformer through passive switches D1 120 and D2 106.

The magnetic flux increases and decreases at rates that are proportionalto the voltage V_(P) 110 on the primary winding. Therefore, when theactive switches S1 104 and S2 116 are on, the magnetic flux increases ata rate substantially proportional to the input voltage V_(IN).Similarly, when the passive switches D1 120 and D2 106 are on, themagnetic flux decreases at a rate substantially proportional to the sumof the input voltage V_(IN) 102 and the reset circuit voltage V_(RC)150.

The transformer T1 114 is typically constructed with a magnetic materialto achieve the desired coupling between primary winding 112 andsecondary winding 134. The magnetic material of the transformer T1 114normally loses desirable properties if the magnetic flux should reach asaturation value. In accordance with the present disclosure, themagnetic flux in the transformer is prevented from reaching itssaturation value.

Although it is possible to measure the magnetic flux directly in thetransformer, methods of doing so are typically not practical for lowcost power supply applications. In contrast, the present disclosure usesa simple indirect technique to indicate the magnitude of the magneticflux. The change in magnetic flux during the time that the activeswitches are on is proportional to the time integral of the voltage onany winding of the transformer. If the initial value of the magneticflux is much less than the saturation value, knowledge of the change inthe magnetic flux during a switching period is normally sufficient toprevent the magnetic flux from reaching its saturation value.

The desired management of magnetic flux can be achieved by integratingthe voltage on a winding of the transformer while the active switchesare on to estimate a peak value of the magnetic flux. Then, the voltageon a winding can be integrated while the active switches are off toensure that magnetic flux decreases by the same amount that it increasedwhile the active switches were on. In the example of FIG. 1, themagnetic flux is managed by measuring and integrating the voltages thatwill be applied to the primary winding 112 of the transformer T1 114.Thus, the disclosed measuring and integrating voltage technique can bedistinguished from a direct measurement of a voltage on a winding of thetransformer.

In the example of FIG. 1, a control circuit 148 receives a feedbacksignal 146 at a feedback terminal 132. Galvanic isolation is typicallymaintained between the input return 108 and the output return 152 in thetransmission of the feedback signal 146 to the feedback terminal 132 byordinary methods known to those skilled in the art, such as for examplethe use of an optical coupler or for example the use of a signaltransformer. The details of the transmission of the feedback signal 146are not discussed in this disclosure to help avoid obscuring theimportant features of the invention.

In the example of FIG. 1, the control circuit 148 receives a firstcurrent I₁ at a reset voltage sensing terminal 120 and a second currentI₂ at line voltage sensing terminal 128. The voltage at the resetvoltage sensing terminal 120 and the voltage at the line voltage sensingterminal 128 are typically low values that are electrically referencedto the input return 108. In one example, the voltages at the resetvoltage sensing terminal 120 and at the line voltage sensing terminal128 are less than approximately three volts whereas the input voltageV_(IN) is typically between 100 volts and 400 volts. Therefore, thecurrent I₂ at the line voltage sensing terminal 128 is substantiallydirectly proportional to the input voltage V_(IN) 102 and inverselyproportional to the value of the resistor R2 126. Similarly, the currentI₁ at the reset voltage sensing terminal 120 is substantially directlyproportional to the sum of the input voltage V_(IN) 102 and the voltageV_(RC) 150 on the reset circuit 118 (and inversely proportional to thevalue of the resistor R1 122). Control circuit 148 responds to thesignals at the reset voltage sensing terminal 120, the line voltagesensing terminal 128, and the feedback terminal 132 to produce a gatesignal 130 that turns the active switches S1 104 and S2 116 on and offto regulate the output voltage V_(O) at the load 144 and to preventsaturation of the transformer T1 114.

FIG. 2 shows details of a typical reset circuit 118 in the two-switchconverter of FIG. 1. The active switches S1 104 and S2 116 turn on for aportion of a switching period. The active switches S1 104 and S2 116 areoff for the remainder of the switching period. The fraction of theswitching period when the active switches are on is often known as theduty ratio. A two-switch forward converter that does not use a resetcircuit 118 has a maximum duty ratio of 50% to ensure that the increasein magnetic flux when the active switches are on is the same as thedecrease in magnetic flux when the active switches are off. That is, theactive switches are not normally closed for more than half the time in acomplete switching period for repetitive switching cycles.

A two-switch forward converter that uses a reset circuit 118 can extendthe maximum duty ratio beyond 50%. The ability to operate at a largerduty ratio has the benefit of permitting operation over a wider range ofinput voltage. Another advantage of the extended duty ratio is thereduction in RMS (root-mean-square) current in the active switches S1104 and S2 116, thereby reducing conduction loss and raising efficiency.

The reset circuit illustrated in the example 200 of FIG. 2 develops avoltage V_(RC) 150 that is substantially constant for several switchingcycles. The reset circuit 118 includes a Zener diode 210, a resistor230, and a capacitor 220. The current from the high-side passive switchD1 120 establishes a voltage V_(RC) 150 on capacitor 220 between thepositive terminal of the input voltage V_(IN) 102 and the high-sidepassive switch D1 110. Zener Diode 210 and resistor 230 substantiallylimit the maximum voltage on capacitor 220.

FIG. 3 is a block diagram 300 that shows several internal functionalblocks of the control circuit 148. The output of the control circuit isthe gate signal 130 that opens and closes the active switches S1 104 andS2 120. In the example of FIG. 1, the active switches S1 104 and S2 116are closed when the signal at the gate terminal 130 is at a high level.In the example of FIG. 1, the active switches S1 104 and S2 116 are openwhen the signal at the gate terminal 130 is at a low level. Anoscillator 310 provides a plurality of timing signals 350, 345, and 360to a pulse width modulator (PWM) 305, saturation prevention circuits315, and logic circuits 320. The pulse width modulator 305 responds to asignal at the feedback terminal 132 to produce a PWMOUT signal 340. ThePWMOUT signal 340 is a timing signal having a switching period T_(S) anda duty ratio required to regulate an output of the power supply.

Saturation prevention circuits 315 receive signals from line voltagesensing terminal 128, from reset voltage sensing terminal 120, and fromgate signal 132 at gate terminal 130. The saturation prevention circuits315 produce signals SOFF 325, DTERM 330, and DENABLE 335 that arereceived by logic circuits 320. Logic circuits 320 process the signalsreceived from the saturation prevention circuits 315 and from the pulsewidth modulator 305 to determine whether or not the PWMOUT signal 340should be allowed to inhibit gate signal 132 to prevent saturation ofthe transformer T1 114. In one example, the SOFF signal 325 demands thatthe active switches stay off. In one example, the DTERM signal 330demands that the active switches turn off immediately. In one example,the DENABLE signal 335 allows the active switches to turn on and off inaccordance with the PWMOUT signal 340.

FIG. 4 is schematic diagram 400 that illustrates the functions ofsaturation prevention circuits included in block 315 of FIG. 3. Acontrolled current source 405 is responsive to the current received byline voltage sensing terminal 128. The value of the controlled currentsource 405 is directly proportional to the current I₂ received at theline voltage sensing terminal 128.

Also in FIG. 4 is a controlled current source 410 that is responsive tothe current received at the reset voltage sensing terminal 120. Thevalue of the controlled current source 410 is directly proportional tothe current I₁ received at the reset voltage sensing terminal 120.

Switches S3 415 and S4 420 open and close in response to a signal at thegate terminal 130. Inverter 425 and AND gate 430 prevent switches S3 415and S4 420 from being closed at the same time.

When the signal at the gate terminal 130 is high, the input voltageV_(IN) 102 is applied to the primary winding 112 of the transformer T1114. The high signal at the gate terminal 130 closes switch S3 415 andopens switch S4 420, allowing current from the controlled current source405 to increase the voltage V_(C) 465 on the integrating capacitor 460.The voltage 465 on the integrating capacitor represents the magneticflux in the transformer T1 114.

When the signal at the gate terminal 130 is low, the sum of the inputvoltage V_(IN) 102 and the voltage V_(RC) 150 from the reset circuit 118is applied to the primary winding 112 of the transformer T1 114. The lowsignal at the gate terminal 130 opens switch S3 415 and closes switch S4420, allowing current from the controlled current source 410 to decreasethe voltage V_(C) 465 on the integrating capacitor 460.

The voltage V_(C) 465 on the integrating capacitor 460 is received bycomparators 435 and 440. Comparator 435 compares the voltage V_(C) 465on the integrating capacitor 460 with an upper threshold voltage V_(TH)470. Comparator 440 compares the voltage V_(C) 465 on the integratingcapacitor 460 with a lower threshold voltage V_(TL) 475. The upperthreshold voltage V_(TH) 470 is greater than the lower threshold voltageV_(TL) 475.

If the voltage V_(C) 465 on the integrating capacitor 460 remains lessthan the upper threshold voltage V_(TH) 470 during the time that thesignal at the gate terminal 130 is high, the signal at the gate terminal130 is allowed to remain high and to go low according to the duty ratiodefined by the PWMOUT signal 340. A low signal at the gate terminal 130opens switch S3 415 and closes switch S4 420 for as long as the voltageV_(C) 465 on the integrating capacitor 460 is above the lower thresholdvoltage V_(TL) 475. When the voltage V_(C) 465 on the integratingcapacitor 460 is no longer greater than the lower threshold voltageV_(TL) 475, the RDIS signal 480 at the output of the comparator 440 goeshigh to open switch S4 420, thereby stopping the discharge of theintegrating capacitor 460.

A DMAX complement signal 455 goes high near the end of every switchingcycle to establish a maximum duty ratio. It is typically necessary todefine a maximum duty ratio to guarantee proper operation of theoscillator 310. FIG. 4 shows that the latch 450 is set by the DMAXcomplement signal 455 when the DMAX compliment signal 455 goes high,thereby setting a high level at the SOFF terminal 325. The logiccircuits 320 force the signal at the gate terminal 130 to be low whenthe signal at the SOFF terminal 325 is high.

The RDIS signal 480 goes high at the end of the discharge of theintegrating capacitor 460. The rising edge of the RDIS signal 480 isdelayed by a leading edge delay 445 to cause DENABLE signal 335 to gohigh after a delay time T_(d). A high level of the DENABLE signal 330when the DMAX complement signal 455 is low resets the latch 450 to bringthe signal at the SOFF terminal 325 low. The RDIS signal 480 goes lowwhen switch S3 415 closes and integrating capacitor 460 charges to raisethe voltage V_(C) 465 above the lower threshold voltage V_(TL) 475. TheDENABLE signal 330 goes low immediately when the RDIS signal 480 goeslow.

If the voltage V_(C) 465 on the integrating capacitor 460 is not lessthan the upper threshold voltage V_(TH) 470, then the output ofcomparator 435 goes to a high level to assert the DTERM signal 330.Logic circuits 320 respond to a high level of the DTERM signal 330 byforcing the signal at the gate terminal 130 to a low level, therebyopening the active switches S1 104 and S2 120 to stop the increase inmagnetic flux in the transformer T1 114.

FIG. 5 is a timing diagram 500 that further illustrates therelationships among the signals in FIG. 3 and FIG. 4 for a conditionthat does not require any action to prevent saturation of thetransformer T1 112. The DMAX complement signal 455 goes low at a time t₀510 that is the beginning of a switching period T_(S) 550. The PWMOUTsignal 340 and the GATE signal 130 go high at time t₁ 515, soon aftertime t₀ 510. The input voltage V_(IN) 102 is applied to the primarywinding 112 of the transformer T1 114 when the GATE signal 130 is high.The voltage V_(C) 420 on the integrating capacitor 460 rises from thelower threshold voltage V_(TL) 475 toward the higher threshold voltageV_(TH) 470 when the GATE signal 130 is high to emulate the increase inmagnetic flux of the transformer T1 114.

The PWMOUT signal 340 and the GATE signal 130 go low at time t₂ 520. Thevoltage V_(C) 420 on the integrating capacitor 460 decreases toward thelower threshold voltage V_(TL) 475 when the GATE signal 130 is low toemulate the decrease in magnetic flux of the transformer T1 114,reaching the lower threshold voltage V_(TL) 475 at time t₃ 525. The RDISsignal 480 goes high at time t₃ 525 to indicate that the magnetic fluxin the transformer T1 114 has returned approximately to the value it hadat time t₁ 515. The DENABLE signal 335 goes high after a delay T_(d) attime t₄ 530.

The SOFF signal 325 goes high when the DMAX complement signal 455 goeshigh at time t₅ 535. The DMAX complement signal 455 goes low at time t₆540 to end the switching period 550 and to start the next switchingperiod 560. When the DMAX complement signal 455 goes low at time t₆ 540,the DENABLE signal 335 resets latch 450 to allow the GATE signal 130 togo high in response to the PWMOUT signal 340. The DTERM signal 330 stayslow in the example of FIG. 5 because the voltage V_(C) 420 on theintegrating capacitor 460 remains less than the higher threshold voltageV_(TH) 470.

FIG. 6 is a timing diagram 600 that further illustrates therelationships among the signals in FIG. 3 and FIG. 4 for a conditionthat requires the circuit to open switches S1 104 and S2 116 to preventsaturation of the transformer T1 112. In the example illustrated in FIG.6, the pulse width modulator 305 has responded to the feedback signal132 to produce a PWMOUT signal 340 that is high from time t₁ 515 untiltime t₇ 560. However, at time t₂ 520 the voltage V_(C) 420 on theintegrating capacitor has reached the higher threshold voltage V_(TH)470, indicating that the magnetic flux in the transformer T1 114 is atits highest desired value. Thus, at time t₂ 520 in FIG. 6 the DTERMsignal 330 goes high, forcing the GATE signal 130 low to open theswitches S1 104 and S2 116, and prevent saturation of the transformer T1114.

FIG. 7 is another timing diagram 700 that further illustrates therelationships among the signals in FIG. 3 and FIG. 4 for a conditionthat directs the circuit to delay the closing of switches S1 104 and S2116 in the next switching period until the magnetic flux in thetransformer T1 114 decreases by at least the amount that it hadincreased during the current switching period. The example illustratedin FIG. 7 shows the voltage V_(C) 420 on integrating capacitor 460 isgreater than the lower threshold voltage V_(TL) 475 at the time t₆ 540that is the end of switching period 550. Time t₆ 540 is also thebeginning of the next switching period 560. PWMOUT signal 340 goes highat time t₈ 705 in switching period 560, but the SOFF signal 325 remainshigh to force the GATE signal 130 low, thereby preventing switches S1104 and S2 116 from closing.

In the example illustrated in FIG. 7, the voltage V_(C) 420 onintegrating capacitor 460 decreases to the lower threshold voltageV_(TL) 475 at time t₉ 710, indicating that the magnetic flux in thetransformer has decreased by approximately the same amount that it hadincreased. The RDIS signal 480 goes high at time t₉ 710. After a delaytime T_(d), the DENABLE signal 335 goes high to reset the latch 450,returning the SOFF signal 325 to a low level at time t₁₀ 715. When theSOFF signal 325 goes low at time t₁₀ 715, the GATE signal 130 is allowedto go high in response to PWMOUT signal 340.

Since the voltage V_(C) 420 on the integrating capacitor 460 is only anindication of the magnetic flux and is not a direct measurement of themagnetic flux, the delay time T_(d) helps compensate for errors in theestimation that might produce a net increase in the magnetic flux. Thedelay time T_(d) provides extra time after the voltage V_(C) on theintegrating capacitor 460 reaches the lower threshold V_(TL) 475 toassure that the magnetic flux has decreased sufficiently to preventsaturation of the transformer T1 114 in subsequent switching periods.

In the next switching period 560, the GATE signal 130 and the PWMOUTsignal 340 go low at time t₁₁ 720. The SOFF signal 325 goes high againat time t₁₂ 725 when the latch 450 is set by the DMAX complement signal455. The end of the next switching period 560 occurs when DMAXcomplement signal 455 goes low at time t₁₃ 730.

FIG. 8, FIG. 9, and FIG. 10 are respective schematic diagrams 800, 900,and 1000 that show an example of an integrated circuit implementation ofthe saturation prevention circuits and logic circuits illustrated inFIG. 3 and FIG. 4. The circuits in FIG. 8 produce the voltage V_(C) 465on the integrating capacitor 460 from the currents I_(I) and I₂ receivedrespectively at the reset voltage sensing terminal 128 and at the linevoltage sensing terminal 120. The example of FIG. 8 uses optional sampleand hold circuits to integrate sampled values of the currents ratherthan to integrate the currents continuously.

A benefit of using sampled values of the currents instead of thecontinuous values of the currents for the integration is to help avoidthe undesirable influence of noise. Thus, the currents received atterminals 120 and 128 are kept as low as possible to help avoiddegrading the efficiency of the power supply since those currents comefrom high voltages. However, keeping the currents low often results inmore susceptibility to corruption from noise. In some applications, theswitching of high voltages and high currents in the power supply couldintroduce noise currents of sufficient magnitude into the terminals 120and 128 to cause a significant error in the value of the voltage V_(C)465 on the integrating capacitor 460. Accordingly, the circuit shown inFIG. 8 samples the currents near the end of each switching period at atime when there is no switching to generate noise. The sampling circuitholds the values of the sampled current for integration throughout thefollowing switching period.

In FIG. 8, line voltage sensing terminal 128 and reset voltage sensingterminal 120 receive respective currents I₂ and I₁. The gates ofP-channel transistors 802 and 818 are held at a voltage V_(BG) 808 toestablish the voltage at terminals 120 and 128. The voltage at terminals120 and 128 is limited to V_(BG) plus a P-channel threshold voltage.N-channel transistors 804 and 810 form a current mirror that limits thesum of the currents in transistors 804 and 810 to the value of currentsource 806. Similarly, N-channel transistors 814 and 820 form a currentminor that limits the sum of the currents in transistors 814 and 820 tothe value of current source 816. The current in P-channel transistor 812is proportional to the current I₁ received at line voltage sensingterminal 128. The current in P-channel transistor 832 is proportional tothe current I₂ received at reset voltage sensing terminal 120.

The voltage between the drain and source terminals of P-channeltransistor 812 is directly proportional to the current I₁. The voltagebetween the drain and source terminals of P-channel transistor 832 isdirectly proportional to the current I₂. A sample signal 838 at therespective gates of P-channel transistors 822 and 834 allows a resetvoltage sampling capacitor 824 and a line voltage sampling capacitor 836to charge to the voltages developed across the P-channel transistors 812and 832 respectively.

The GATE complement signal 830 at the gate of P-channel transistor 840applies the line sense voltage V_(LS) of line voltage sense capacitor836 between the source and gate of P-channel transistor 842 to produce acurrent K₂I₂ that charges integrating capacitor 460 while the switchesS1 104 and S2 116 are closed. Thus, the integrating capacitor 460charges with a current that is directly proportional to the inputvoltage V_(IN) 102. Similarly, the reset sense voltage V_(RS) of resetvoltage sense capacitor 824 is applied between the source and gate ofP-channel transistor 826 to produce a current in P-channel transistor826 and N-channel transistor 828 that is mirrored by N-channeltransistor 844 to produce a current K₁I₁ that discharges integratingcapacitor 460 when N-channel transistors 846 and 848 are both on. Thus,the integrating capacitor 460 discharges with a current that is directlyproportional to the reset voltage of the transformer T1 114.

The RDIS complement signal 880 at the gate of N-channel transistor 846prevents the discharge of the integrating capacitor 460 when the voltageV_(C) 465 on integrating capacitor 460 is below a lower thresholdvoltage V_(TL) 475. The GATE complement signal 830 at the gate ofN-channel transistor 848 prevents the discharge of the integratingcapacitor 460 when the switches S1 104 and S2 116 are closed.

FIG. 9 shows that the RDIS complement signal 880 is the output ofcomparator 440 that determines whether or not the voltage V_(C) 465 onthe integrating capacitor 460 is greater than the lower thresholdvoltage V_(TL) 475. Current source 905, P-channel transistor 910,N-channel transistor 915, timing capacitor 920, and inverter 925, andinverter 950 perform the function of the leading edge delay circuit 445.

NAND gates 955 and 960 perform the function of the latch 450. Inverter940 receives the DMAX complement signal 455 to produce the DMAX signal995 at one input of NAND gate 960. The latch is set when the DMAX signal995 goes low. The output of an inverter 925 is the DENABLE complementsignal 935. The latch formed by NAND gates 955 and 960 is reset when theDENABLE complement signal 935 goes low.

Inverter 945 produces a DTERM complement signal 930 from the output ofcomparator 435. Inverter 950 produces a DENABLE signal 335 from theDENABLE complement signal 935.

FIG. 10 illustrates a circuit 1000 that shows how GATE signal 130 andGATE complement signal 830 are derived from the DMAX complement signal455. The NOR gate 1025 receives the DMAX complement signal 455 at oneinput. The other input of the NOR gate 1025 receives the DMAX complementsignal 455 after it is delayed by inverters 1005, 1010, and 1015. Theoutput 1040 of NOR gate 1025 is a train of pulses that sets a latch 1035at the start of each switching period. The latch 1035 is reset when theoutput of NAND gate 1030 goes high. OR gate 1045 is used to gate the Qcomplement output of latch 1035 with the SOFF signal 325. The output ofOR gate 1045 is GATE complement signal 830. Inverter 1020 inverts theGATE complement signal 830 to produce the GATE signal 130.

FIG. 11 is a timing diagram 1100 that shows signals in the exampleintegrated circuit implementation of FIG. 8, FIG. 9, and FIG. 10 forsimilar conditions as presented in FIG. 5. FIG. 11 shows the SAMPLEcomplement signal 830 along with the complements of several signalsshown in FIG. 5. FIG. 11 shows that the SAMPLE complement signal 830goes low just before the beginning of the switching period at times t₀550 and t₆ 540.

FIG. 12 is a timing diagram 1200 that shows signals in the exampleintegrated circuit implementation of FIG. 8, FIG. 9, and FIG. 10 forsimilar conditions as presented in FIG. 6.

FIG. 13 is a timing diagram 1300 that shows signals in the exampleintegrated circuit implementation of FIG. 8, FIG. 9, and FIG. 10 forsimilar conditions as presented in FIG. 7.

FIG. 14 is a flow diagram 1400 that illustrates a method of theinvention. After starting in block 1405, active switches are closed inblock 1410. Integration of the line voltage begins in block 1415 afterthe active switches close in block 1410. Decision block 1420 comparesthe integrated line voltage to a threshold value K_(TH). If the integralof the line voltage is less than the threshold value K_(TH), then thepulse width modulator is allowed to open the active switches in block1430. If the integral of the line voltage is not less than the thresholdvalue K_(TH), then the active switches are opened in block 1425. Thereset voltage for the transformer is integrated in block 1435 after theactive switches open.

Decision block 1440 compares the difference between the value of theintegrated input voltage and the value of the integrated reset voltageto a lower threshold value K_(TL) during the integration of the resetvoltage. The integration of the reset voltage continues as long as thedifference between the integrated input voltage and the integrated resetvoltage is greater than the lower threshold value K. The integrationstops in block 1445 when the difference between the integrated inputvoltage and the integrated reset voltage is not greater than the lowerthreshold value K.

After a delay time T_(d) in block 1450, the pulse width modulator isallowed to close the active switches in block 1455. The flow continuesin block 1410.

The invention is not limited to the example two-switch forwardconverter, but can be applied easily to a single-switch forwardconverter as illustrated in FIG. 15 and FIG. 16. FIG. 15 is schematicdiagram 1500 that shows the salient features of a single-switch forwardconverter that uses many of the same components as the two-switchforward converter of FIG. 1. The control circuit 1548 for thesingle-switch forward converter is modified from the control circuit 148for the two-switch forward converter as illustrated in FIG. 16.

FIG. 16 is schematic diagram 1600 that shows that the controlled currentsource 1610 that discharges the integration capacitor 460 has adifferent value in the application to the single-switch forwardconverter than the controlled current source 410 has in the applicationto the two-switch forward converter. In the application to thesingle-switch forward converter, the value of controlled current source1610 is the difference (K₃I₁- K₄I₂) between a value directlyproportional to the current in resistor R2 126 and a value directlyproportional to the current in resistor R1 122, whereas the value ofcontrolled current source 410 is directly proportional to the current inresistor R1 122 only.

The above description of illustrated examples of the present invention,including what is described in the Abstract, are not intended to beexhaustive or to be limiting as to the precise forms disclosed. Whilespecific embodiments of, and examples for, the invention are describedherein for illustrative purposes, various equivalent modifications arepossible without departing from the broader spirit and scope of thepresent invention. Indeed, it is appreciated that the specific voltages,currents, frequencies, power range values, times, etc., are provided forexplanation purposes and that other values may also be employed in otherembodiments and examples in accordance with the teachings of the presentinvention.

These modifications can be made to examples of the invention in light ofthe above detailed description. The terms used in the following claimsshould not be construed to limit the invention to the specificembodiments disclosed in the specification and the claims. Rather, thescope is to be determined entirely by the following claims, which are tobe construed in accordance with established doctrines of claiminterpretation. The present specification and figures are accordingly tobe regarded as illustrative rather than restrictive.

1. A control circuit for use in a power converter, the control circuitcomprising: a pulse width modulator coupled to generate a pulse widthmodulation signal in response to a feedback signal; a logic circuitcoupled to turn a switch on and off to regulate an output of the powerconverter in response to the pulse width modulation signal; and asaturation prevention circuit coupled to the logic circuit to assert afirst signal when a first integral value of an input voltage of thepower converter reaches a first threshold value and to assert a secondsignal after a delay time that begins when a difference between thefirst integral value and a second integral value of a reset voltage of atransformer of the power converter falls to a second threshold value,wherein the logic circuits are adapted to turn off the switch when thefirst signal is asserted, and to allow the switch to turn on and off inaccordance with the pulse width modulation signal when the second signalis asserted.
 2. The control circuit of claim 1, further comprising: aline voltage sensing terminal coupled to receive a first current that isdirectly proportional to the input voltage; and a reset voltage sensingterminal coupled to receive a second current that is directlyproportional to the reset voltage.
 3. The control circuit of claim 2,wherein the saturation prevention circuit further comprises: a firstcontrolled current source coupled to produce a charging current that isproportional to the first current; a second controlled current sourcecoupled to produce a discharging current that is proportional to thesecond current; and an integrating capacitor coupled to be charged withthe charging current while the switch is on and to be discharged withthe discharging current when the switch is off, wherein a voltage on theintegrating capacitor is representative of a magnetic flux of thetransformer, and wherein the saturation prevention circuit is adapted toassert the first signal when the voltage on the integrating capacitor isgreater than or equal to the first threshold value, and to assert thesecond signal after the delay time that begins when the integratingcapacitor discharges to the second threshold value.
 4. The controlcircuit of claim 3, wherein the second controlled current source iscoupled to begin discharging the integrating capacitor in response tothe logic circuit turning off the switch.
 5. The control circuit ofclaim 3, wherein the second controlled current source is coupled todischarge the integrating capacitor if the switch is turned off and ifthe voltage on the integrating capacitor is greater than or equal to thesecond threshold value.
 6. The control circuit of claim 3, wherein thefirst controlled current source is coupled to begin charging theintegrating capacitor in response to the logic circuit turning on theswitch.
 7. The control circuit of claim 6, wherein the first controlledcurrent source is coupled to end charging the integrating capacitor inresponse to the logic circuits turning off the switch.
 8. The controlcircuit of claim 3, wherein the saturation prevention circuit furthercomprises: a first comparator coupled to compare the voltage on theintegrating capacitor with the first threshold value and to output thefirst signal; and a second comparator coupled to compare the voltage onthe integrating capacitor with the second threshold value; and a leadingedge delay circuit coupled to an output of the second comparator and togenerate the second signal.
 9. The control circuit of claim 3, whereinthe saturation prevention circuit further comprises a latch coupled togenerate a third signal to keep the switch turned off when the latch isset, wherein the latch is set in response to a maximum duty ratio signalbeing asserted and is reset in response to the voltage on theintegrating capacitor discharging to the second threshold value.